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SC4501 2Amp, 2MHz Step-Up Switching Regulator with Soft-Start
POWER MANAGEMENT Description
The SC4501 is a high-frequency current-mode step-up switching regulator with an integrated 2A power transistor. Its high switching frequency (programmable up to 2MHz) allows the use of tiny surface-mount external passive components. Programmable soft-start eliminates high inrush current during start-up. The internal switch is rated at 32V making the converter suitable for high voltage applications such as Boost, SEPIC and Flyback. The operating frequency of the SC4501 can be set with an external resistor. The ability to set the operating frequency gives the SC4501 design flexibilities. A dedicated COMP pin allows optimization of the loop response. The SC4501 is available in thermally enhanced 8-Pin MSOP and 10-pin MLPD packages.
Features
u u u u u u u u u Low saturation voltage switch: 220mV at 2A Constant switching frequency current-mode control Programmable switching frequency up to 2MHz Soft-Start function Input voltage range from 1.4V to 16V Output voltage up to 32V Low shutdown current Adjustable undervoltage lockout threshold Small low-profile thermally enhanced packages
Applications
u u u u u u u
Flat screen LCD bias supplies TFT bias supplies XDSL power supplies Medical equipment Digital video cameras Portables devices White LED power supplies
Typical Application Circuit
VIN 5V 6 OFF ON 3 C1 2.2F 8 IN SHDN SC4501 SS GND C3 47nF 4 COMP ROSC 7 R4 C4 1 R3 C6 R2 20K 5 SW FB 2 C2 10F L1 D1 VOUT 12V, 0.7A R1 174K
10BQ015
Efficiency
95 10.5H, 700KHz 90 85 4.7H, 1.4MHz
Efficiency (%)
80 3.3H, 2MHz 75 70 65 60
All Capacitors are Ceramic. MSOP-8 Pinout
f / MHz 0.7 1.35 2 R3 / K 22.1 30.9 63.4 R4 / K 22.1 9.31 4.75 C4 / pF 2200 820 470 C6 / pF 22 L1 / H 10.5 (Falco D08019) 4.7 (Falco D08017) 3.3 (Coilcraft DO1813P)
55 50 0.0 0.1 0.2 0.3 0.4
VIN = 5V VOUT = 12V 0.5 0.6 0.7
Load Current (A)
Figure 1(a). 5V to 12V Boost Converter.
Figure 1(b). Efficiencies of 5V to 12V Boost Converters at 700KHz, 1.4MHz and 2MHz.
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Revision: October 25, 2005
SC4501
POWER MANAGEMENT Absolute Maximum Rating
Exceeding the specifications below may result in permanent damage to the device, or device malfunction. Operation outside of the parameters specified in the Electrical Characteristics section is not implied.
Parameter Supply Voltage SW Voltage FB Voltages SHDN Voltage Operating Temperature Range Thermal Resistance Junction to Ambient (MSOP-8) Thermal Resistance Junction to Ambient (MLPD-10) Maximum Junction Temperature Storage Temperature Range Lead Temperature (Soldering)10 sec ESD Rating (Human Body Model)
Symbol VIN VSW VFB VSHDN TA JA JA TJ TSTG TLEAD ESD
Typ -0.3 to 18 -0.3 to 32 -0.3 to 2.5 -0.3 to VIN + 1 -40 to +85 40
Units V V V V C C/W
40 160 -65 to +150 260 2000 C C C V
Electrical Characteristics
Unless specified: VIN = 2V, SHDN = 1.5V, ROSC = 7.68k, -40C < T A = TJ < 85C
Parameter Undervoltage Lockout Threshold Maximum Operating Voltage Feedback Voltage Feedback Voltage Line Regulation FB Pin Bias Current Error Amplifier Transconductance Error Amplifier Open-Loop Gain COMP Source Current COMP Sink Current VIN Quiescent Supply Current VIN Supply Current in Shutdown Switching Frequency Maximum Duty Cycle Minimum Duty Cycle Switch Current Limit Switch Saturation Voltage
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Test Conditions
Min
Typ 1.3
Max 1.4 16
Unit V V V V %
TA = 25C -40C < TA < 85C 1.5V < VIN < 16V
1.224 1.217
1.242
1.260 1.267
0.01 40 60 49 80
nA -1 dB A A
VFB = 1.1V VFB = 1.4V VSHDN = 1.5V, VCOMP = 0 ( Not Switching ) VSHDN = 0 1.3 85
5 5 1.1 10 1.5 90 0 2 2.8 220 350 1.6 18 1.7
mA A MHz % % A mV
ISW = 2A
2
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SC4501
POWER MANAGEMENT Electrical Characteristics (Cont.)
Unless specified: VIN = 2V, SHDN = 1.5V, ROSC = 7.68k, -40C < T A = TJ < 85C
Parameter Switch Leakage Current Shutdown Threshold Voltage Shutdown Pin Current Soft-Start Charging Current Thermal Shutdown Temperature Thermal Shutdown Hysteresis
Test Conditions VSW = 5V
Min
Typ 0.01
Max 1 1.18
Unit A V A
1.02 VSHDN = 1.2V VSHDN = 0 VSS = 0.3V
1.1 -4.6 0 1.5 160 10
0.1
A A C C
Pin Configurations
TOP VIEW
Ordering Information
Device(1)(2) SC4501MLTRT SC4501MSETRT(3) SC4501EVB Package MLPD-10 -40 to 85C MSOP-8-EDP Evaluation Board Temp. Range( TA)
(10 Pin - MLPD, 3 x 3mm)
Notes: (1) Only available in tape and reel packaging. A reel contains 3000 devices for MLP package and 2500 devices for MSOP. (2) Lead free product. This product is fully WEEE and RoHS compliant. (3) Contact factory for availability.
TOP VIEW COMP 1 FB 2 SHDN 3 GND 4 8 SS 7 ROSC 6 IN 5 SW
(8 Pin MSOP-EDP)
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SC4501
POWER MANAGEMENT Pin Descriptions (MSOP-8) Block Diagram
Pin 1 2 Pin Name COMP FB Pin Function The output of the internal transconductance error amplifier. This pin is used for loop compensation. The inverting input of the error amplifier. Tie to an external resistive divider to set the output voltage. Shutdown Pin. The accurate 1.1V shutdown threshold and the 4.6uA shutdown pin current hysteresis allow the user to set the undervoltage lockout threshold and hysteresis for the switching regulator. Pulling this pin below 0.1V causes the converter to shut down to low quiescent current. Tie this pin to IN if the UVLO and the shutdown features are not used. This pin should not be left floating. Ground. Tie to the ground plane. Collector of the internal power transistor. Connect to the boost inductor and the rectifying diode. Power Supply Pin. Bypassed with capacitors close to the pin. A resistor from this pin to the ground sets the switching frequency. Soft-Start Pin. A capacitor from this pin to the ground lengthens the start-up time and reduces startup current. The exposed pad must be soldered to the ground plane on the PCB for good thermal conduction.
3
SHDN
4 5 6 7 8
GND SW IN ROSC SS Exposed Pad
Block Diagram
IN 6 4.6A SW 5
SHDN 3
+
CMP
1.1V VOLTAGE REFERENCE THERMAL SHUTDOW N ENABLE CLK
INTERNAL SUPPLY
REG
1.242V FB 2 COMP 1 SS 8
+
-
EA REG 1.5A
PWM
R Q S
+
+
ILIM I-LIMIT
R SENSE
REG_GOOD ENABLE
+ +
+
ISEN
4 GND
ROSC 7
CLK
OSCILLATOR
SLOP E COMP
Figure 2. SC4501 (MSOP-8) Block Diagram.
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SC4501
POWER MANAGEMENT Pin Descriptions (MLPD - 10)
Pin 1 2 Pin Name COMP FB Pin Function The output of the internal transconductance error amplifier. This pin is used for loop compensation. The inverting input of the error amplifier. Tie to an external resistive divider to set the output voltage. Shutdown Pin. The accurate 1.1V shutdown threshold and the 4.6uA shutdown pin current hysteresis allow the user to set the undervoltage lockout threshold and hysteresis for the switching regulator. Pulling this pin below 0.1V causes the converter to shut down to low quiescent current. Tie this pin to IN if the UVLO and the shutdown features are not used. This pin should not be left floating. Ground. Tie both pins to the ground plane. Pins 4 and 5 are not internally connected. Collector of the internal power transistor. Connect to the boost inductor and the rectifying diode. Power Supply Pin. Bypassed with capacitors close to the pin. A resistor from this pin to the ground sets the switching frequency. Soft-Start Pin. A capacitor from this pin to the ground lengthens the start-up time and reduces startup current. The exposed pad must be soldered to the ground plane on the PCB for good thermal conduction.
3
SHDN
4,5 6,7 8 9 10
GND SW IN ROSC SS Exposed Pad
Block Diagram
IN 8 4.6A SW SW 6 7
SHDN 3
+
CMP
1.1V VOLTAGE REFERENC E THERMAL SHUTDOWN ENABL E CLK
INTERNAL SUPPLY
REG
1.242V FB 2 COMP 1 SS 10 REG_GOOD ENABL E
+
-
EA REG 1.5A
PWM
R S Q
+
+
ILIM I-LIMIT
RSENSE
+ +
+
ISEN
4 5 GND GND
ROSC 9
CLK
OSCILLATOR
SLOPE COMP
Figure 3. SC4501 (MLPD-10) Block Diagram.
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SC4501
POWER MANAGEMENT Typical Characteristics
Feedback Voltage vs Temperature
1.3
100
ROSC vs Switching Frequency
1.7
Switching Frequency vs Temperature
ROSC = 7.68K
Feedback Voltage (V)
1.25
VIN = 2V ROSC (K ) 25C
10
Frequency (MHz)
1.6
VIN = 12V
1.5
VIN = 2V
1.2
1.4
1.15 -50 -25 0 25 50 75 100 125
1 0.0 0.5 1.0 1.5 2.0 2.5 3.0
1.3 -50 -25 0 25 50 75 100 125
Temperature (C)
Frequency (MHz)
Temperature (C)
Switch Saturation Voltage vs Switch Current
400
3
Switch Current Limit vs Temperature
1.5
Minimum VIN vs Temperature
VCESAT (mV)
85C
200
2.6
Input Voltage (V)
-50 -25 0 25 50 75 100
Current Limit (A)
300
25C
2.8
1.4
1.3
-40C
100
2.4
1.2
2.2
1.1
0 0 0.5 1 1.5 2 2.5 3
2
1 -50 -25 0 25 50 75 100 125
Switch Current (A)
Temperature (C)
Temperature (C)
VIN Quiescent Current vs Temperature
1.3
VIN Current in Shutdown vs Input Voltage
50
1.20
Shutdown Threshold vs Temperature
VIN = 2V
Not Switching
1.2
40
Shutdown Threshold (V)
VSHDN = 0
VIN Current (mA)
1.1
VIN Current ( A)
VIN = 16V
-40C
30
1.15
125C
20
1.10
1
VIN = 2V
0.9
1.05
10
0.8 -50 -25 0 25 50 75 100 125
0 0 5 10 15 20
1.00 -50 -25 0 25 50 75 100 125
Temperature (C)
Input Voltage (V)
Temperature (C)
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SC4501
POWER MANAGEMENT Typical Characteristics
VIN Current vs SHDN Pin Voltage
1.2 VIN = 2V 1
0.08 0.1 VIN = 2V
VIN Current vs SHDN Pin Voltage
-3
Shutdown Pin Current vs Temperature
V SHDN = 1.25V
VIN Current (mA)
VIN Current (mA)
0.06
Current (A)
0.8 0.6 0.4 0.2 0 0 0.5 1
125C
25C
-4
VIN = 2V
0.04
-5
VIN = 12V
-40C
125C -40C
0.02
0
-6
1.5
0
0.2
0.4
0.6
0.8
1
1.2
-50
-25
0
25
50
75
100
125
SHDN Voltage (V)
SHDN Voltage (V)
Temperature (C)
Soft-Start Charging Current vs Temperature
2
Transconductance vs Temperature
80
V SS = 0.3V
VIN = 2V Transconductance ( )
70
-1
1.8
Current (A)
1.6
60
1.4
50
1.2
40
1 -50 -25 0 25 50 75 100 125
30 -50 -25 0 25 50 75 100 125
Temperature (C)
Temperature (C)
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SC4501
POWER MANAGEMENT Operation
The SC4501 is a programmable constant-frequency peak current-mode step-up switching regulator with an integrated 2A power transistor. Referring to the block diagrams in Figures 2 and 3, the power transistor is switched on at the trailing edge of the clock. Switch current is sensed with an integrated sense resistor. The sensed current is summed with the slope-compensating ramp before compared to the output of the error amplifier EA. The PWM comparator trip point determines the switch turn-on pulse width. The current-limit comparator ILIM turns off the power switch when the switch current exceeds the 2.8A current-limit threshold. ILIM therefore provides cycle-by-cycle current limit. Current-limit is not affected by slope compensation because the current comparator ILIM is not in the PWM signal path. Current-mode switching regulators utilize a dual-loop feedback control system. In the SC4501 the amplifier output COMP controls the peak inductor current. This is the inner current loop. The double reactive poles of the output LC filter are reduced to a single real pole by the inner current loop, easing loop compensation. Fast transient response can be obtained with a simple Type-2 compensation network. In the outer loop, the error amplifier regulates the output voltage. The switching frequency of the SC4501 can be programmed up to 2MHz with an external resistor from the ROSC pin to the ground. For converters requiring extreme duty cycles, the operating frequency can be lowered to maintain the necessary minimum on time or the minimum off time. The SC4501 requires a minimum input of 1.4V to operate. A voltage higher than 1.1V at the shutdown pin enables the internal linear regulator REG in the SC4501. After VREG becomes valid, the soft-start capacitor is charged with a 1.5A current source. A PNP transistor clamps the output of the error amplifier as the soft-start capacitor voltage rises. Since the COMP voltage controls the peak inductor current, the inductor current is ramped gradually during soft-start, preventing high input start-up current. Under fault conditions (VIN<1.4V or over temperature) or when the shutdown pin is pulled below 1.1V, the soft-start capacitor is discharged to ground. Pulling the shutdown pin below 0.1V reduces the total supply current to 10A.
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Application Information
Setting the Output Voltage An external resistive divider R1 and R2 with its center tap tied to the FB pin (Figure 4) sets the output voltage.
V R1 = R2 OUT - 1 1.242V
VOUT
(1)
R1 40nA 2
SC4501
FB
R2
Figure 4. The Output Voltage is set with a Resistive Divider
The input bias current of the error amplifier will introduce an error of:
VOUT 40nA (R1 // R2 )100 = % VOUT 1.242V
(2)
The percentage error of a VOUT = 5V converter with R1 = 100K and R2 = 301K is
VOUT 40nA (100K // 301K )100 = = 0.24% VOUT 1.242V
Operating Frequency and Efficiency Switching frequency of SC4501 is set with an external resistor from the ROSC pin to the ground. A graph showing the relationship between ROSC and switching frequency is given in the "Typical Characteristics". High frequency operation reduces the size of passive components but switching losses are higher. The efficiencies of 5V to 12V converters operating at 700KHz, 1.35MHz and 2MHz are shown in Figure 1(b). The peak efficiency of the SC4501 appears to decrease 0.5% for every 100KHz increase in frequency.
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SC4501
POWER MANAGEMENT Application Information
Duty Cycle The duty cycle D of a boost converter is:
VIN VOUT + VD D= V 1 - CESAT VOUT + VD 1-
It is worth noting that IOUTMAX is directly proportional to the
VIN ratio V . Equation (4) over-estimates the maximum OUT
output current at high frequencies (>1MHz) since switching losses are neglected in its derivation. Nevertheless it is a useful first-order approximation. Using VCESAT = 0.3V, VD = 0.5V and ILIM = 2A in (3) and (4), the maximum output currents for three VIN and VOUT combinations are shown in Table 1.
(3)
where VCESAT is the switch saturation voltage and VD is voltage drop across the rectifying diode. Maximum Output Current
VIN ( V )
VOUT ( V ) 12 5 12
D 0.820 0.423 0.615
IOUTMAX ( A ) 0.35 1.14 0.76
In a boost switching regulator the inductor is connected to the input. The DC inductor current is the input current. When the power switch is turned on, the inductor current flows into the switch. When the power switch is off, the inductor current flows through the rectifying diode to the output. The output current is the average diode current. The diode current waveform is trapezoidal with pulse width (1 - D)T (Figure 5). The output current available from a boost converter therefore depends on the converter operating duty cycle. The power switch current in the SC4501 is internally limited to 2A. This is also the maximum inductor or the input current. By estimating the conduction losses in both the switch and the diode, an expression of the maximum available output current of a boost converter can be derived:
ILIM VIN D VD - D(VD - VCESAT ) 1 - 45 - VOUT VIN
2.5 3.3 5
Table 1. Calculated Maximum Output Current [ Equation (4)]
Considerations for High Frequency Operation The operating duty cycle of a boost converter decreases as VIN approaches VOUT. The PWM modulating ramp in a current-mode switching regulator is the sensed current ramp of the control switch. This current ramp is absent unless the switch is turned on. The intersection of this ramp with the output of the voltage feedback error amplifier determines the switch pulse width. The propagation delay time required to immediately turn off the switch after it is turned on is the minimum switch on time. Regulator closed-loop measurement shows that the SC4501 has a minimum on time of about 150ns at room temperature. The power switch in the SC4501 is either not turned on at all or for at least 150ns. If the required switch on time is shorter than the minimum on time, the regulator will either skip cycles or it will start to jitter. Example: Determine the maximum operating frequency of a Li-ion cell to 5V converter using the SC4501. Assuming that VD=0.5V, VCESAT=0.3V and VIN=2.6 - 4.2V, the minimum duty ratio can be found using (3).
DMIN 4.2 5 + 0.5 = 0.25 = 0.3 1- 5 + 0 .5 1-
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IOUTMAX =
(4)
where ILIM is the switch current limit.
IIN Inductor Current ON OFF ON Switch Current
Diode Current
DT ON OFF
(1-D)T IOUT ON OFF ON
Figure 5. Current Waveforms in a Boost Regulator
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SC4501
POWER MANAGEMENT Application Information
The absolute maximum operating frequency of the
DMIN 0.25 = = 1.67MHz . The 150ns 150ns actual operating frequency needs to be lower to allow for modulating headroom.
converter is therefore The power transistor in the SC4501 is turned off every switching period for an interval determined by the discharge time of the oscillator ramp and the propagation delay of the power switch. This minimum off time limits the maximum duty cycle of the regulator at a given
D(VIN - VCESAT ) (5) fL where f is the switching frequency and L is the inductance. IL =
Substituting (3) into (5) and neglecting VCESAT ,
IL = VIN VIN 1 - fL VOUT + VD
(6)
VOUT switching frequency. A boost converter with high V ratio In
requires long switch on time and high duty cycle. If the required duty cycle is higher than the attainable maximum, then the converter will operate in dropout. (Dropout is a condition in which the regulator cannot attain its set output voltage below current limit.) The minimum off times of closed-loop boost converters set to various output voltages were measured by lowering their input voltages until dropout occurs. It was found that the minimum off time of the SC4501 ranged from 80 to 110ns at room temperature. Beware of dropout when operating at very low input voltages (1.5-2V) and with off times approaching 110ns. Shorten the PCB trace between the power source and the device input pin, as line drop may be a significant percentage of the input voltage. A regulator in dropout may appear as if it is in current limit. The cycle-by-cycle current limit of the SC4501 is duty-cycle and input voltage invariant and is typically 2.8A. If the switch current limit is not at least 2A, then the converter is likely in dropout. The switching frequency should then be lowered to improve controllability. Both the minimum on time and the minimum off time reduce control range of the PWM regulator. Bench measurement showed that reduced modulating range started to be a problem at frequencies over 2MHz. Although the oscillator is capable of running well above 2MHz, controllability limits the maximum operating frequency. Inductor Selection The inductor ripple current I L of a boost converter operating in continuous-conduction mode is
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In current-mode control, the slope of the modulating (sensed switch current) ramp should be steep enough to lessen jittery tendency but not so steep that large flux swing decreases efficiency. Inductor ripple current IL between 25-40% of the peak inductor current limit is a good compromise. Inductors so chosen are optimized in size and DCR. Setting IL = 0.3*(2) = 0.6A, VD=0.5V in (6),
L= VIN fIL VIN V VIN 1 - = IN 1 - 0. 6 f VOUT + VD VOUT + 0.5
(7)
where L is in H and f is in MHz. Equation (6) shows that for a given VOUT, IL is the highest when VIN = . If VIN varies over a wide range, then 2 choose L based on the nominal input voltage. The saturation current of the inductor should be 20-30% higher than the peak current limit (2.8A). Low-cost powder iron cores are not suitable for high-frequency switching power supplies due to their high core losses. Inductors with ferrite cores should be used. Input Capacitor The input current in a boost converter is the inductor current, which is continuous with low RMS current ripples. A 2.2-4.7F ceramic input capacitor is adequate for most applications. Output Capacitor Both ceramic and low ESR tantalum capacitors can be used as output filtering capacitors. Multi-layer ceramic capacitors, due to their extremely low ESR (<5m), are the best choice. Use ceramic capacitors with stable temperature and voltage characteristics. One may be tempted to use Z5U and Y5V ceramic capacitors for output filtering because of their high capacitance and
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(VOUT + VD )
SC4501
POWER MANAGEMENT Application Information
small sizes. However these types of capacitors have high temperature and high voltage coefficients. For example, the capacitance of a Z5U capacitor can drop below 60% of its room temperature value at -25C and 90C. X5R ceramic capacitors, which have stable temperature and voltage coefficients, are the preferred type. The diode current waveform in Figure 5 is discontinuous with high ripple-content. In a buck converter the inductor ripple current IL determines the output ripple voltage. The output ripple voltage of a boost regulator is however much higher and is determined by the absolute inductor current. Decreasing the inductor ripple current does not appreciably reduce the output ripple voltage. The current flowing in the output filter capacitor is the difference between the diode current and the output current. This capacitor current has a RMS value of: forward voltages). This is because the diode conduction interval is much longer than that of the transistor. Converter efficiency will be improved if the voltage drop across the diode is lower. The rectifying diodes should be placed close to the SW pins of the SC4501 to minimize ringing due to trace inductance. Surface-mount equivalents of 1N5817, 1N5818, MBRM120 (ON Semi) and 10BQ015 (IRF) are all suitable. Soft-Start Soft-start prevents a DC-DC converter from drawing excessive current (equal to the switch current limit) from the power source during start up. If the soft-start time is made sufficiently long, then the output will enter regulation without overshoot. An external capacitor from the SS pin to the ground and an internal 1.5A charging current source set the soft-start time. The soft-start voltage ramp at the SS pin clamps the error amplifier output. During regulator start-up, COMP voltage follows the SS voltage. The converter starts to switch when its COMP voltage exceeds 0.7V. The peak inductor current is gradually increased until the converter output comes into regulation. If the shutdown pin is forced below 1.1V or if fault is detected, then the soft-start capacitor will be discharged to ground immediately. The SS pin can be left open if soft-start is not required. Shutdown The input voltage and shutdown pin voltage must be greater than 1.4V and 1.1V respectively to enable the SC4501. Forcing the shutdown pin below 1.1V stops switching. Pulling this pin below 0.1V completely shuts off the SC4501. The total VIN current decreases to 10A at 2V. Figure 6 shows several ways of interfacing the control logic to the shutdown pin. Beware that the shutdown pin is a high impedance pin. It should always be driven from a lowimpedance source or tied to a resistive divider. Floating the shutdown pin will result in undefined voltage. In Figure 6(c) the shutdown pin is driven from a logic gate whose VOH is higher than the supply voltage of the SC4501. The diode clamps the maximum shutdown pin voltage to one diode voltage above the input power supply.
IOUT
VOUT -1 VIN
(8)
If a tantalum capacitor is used, then its ripple current rating in addition to its ESR will need to be considered. When the switch is turned on, the output capacitor supplies the load current IOUT (Figure 5). The output ripple voltage due to charging and discharging of the output capacitor is therefore:
VOUT =
IOUTDT COUT
(9)
For most applications, a 10-22F ceramic capacitor is sufficient for output filtering. It is worth noting that the output ripple voltage due to discharging of a 10F ceramic capacitor (9) is higher than that due to its ESR. Rectifying Diode For high efficiency, Schottky barrier diodes should be used as rectifying diodes for the SC4501. These diodes should have a RMS current rating of at least 1A and a reverse blocking voltage of at least a few Volts higher than the output voltage. For switching regulators operating at low duty cycles (i.e. low output voltage to input voltage conversion ratios), it is beneficial to use rectifying diodes with somewhat higher RMS current ratings (thus lower
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SC4501
POWER MANAGEMENT Application Information
IN SC4501
IN SC4501
SHDN
SHDN
(a)
(b)
VIN 1N4148
IN SC4501
IN SC4501
SHDN
SHDN
(c)
(d)
Figure 6. Methods of Driving the Shutdown Pin (a) Directly Driven from a Logic Gate (b) Driven from an Open-drain N-channel MOSFET or an Open-Collector NPN Transistor (VOL < 0.1V) (c) Driven from a Logic Gate with VOH > VIN (d) Combining Shutdown with Programmed UVLO (See Section Below).
Programming Undervoltage Lockout
The SC4501 has an internal VIN undervoltage lockout (UVLO) threshold of 1.4V. The transition from idle to switching is abrupt but there is no hysteresis. If the input voltage ramp rate is slow and the input bypass is limited, then sudden turn on of the power transistor will cause a dip in the line voltage. Switching will stop if VIN falls below the internal UVLO threshold. The resulting output voltage rise may be non-monotonic. The 1.1V disable threshold of the SC4501 can be used in conjunction with a resistive voltage divider to raise the UVLO threshold and to add an UVLO hysteresis. Figure 7 shows the scheme. Both VH and VL (the desired upper and the lower UVLO threshold voltages) are determined by the 1.1V threshold crossings,
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VH and VL are therefore:
R VH = 1 + 3 (1.1 V ) R4 VL = VH - VHYS = VH - IHYSR3
(10)
Re-arranging,
R3 =
R4 =
12
VHYS IHYS
R3 VH -1 1 .1
(11)
(12)
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SC4501
POWER MANAGEMENT Application Information
The turn off voltage is:
VL = VH - VHYS = 2.75 - 0.69 = 2.06 V > 1.4 V .
IN 6/8
Frequency Compensation
I HYS 4.6A
R3
SWITCH CLOSED WHEN Y = "1" SHDN 3 + Y COMPARATOR
Figure 8 shows the equivalent circuit of a boost converter using the SC4501. The output filter capacitor and the load form an output pole at frequency:
p2 = -
2IOUT 2 =- VOUTC2 ROUTC2
(13)
R4
1.1V
where C2 is the output capacitor and ROUT = equivalent load resistance.
VOUT is the IOUT
SC4501
Figure 7. Programmable Hysteretic UVLO Circuit
The zero formed by C2 and its equivalent series resistance (ESR) is neglected due to low ESR of the ceramic output capacitor. There is also a right half plane (RHP) zero at angular frequency:
Z 2 = ROUT (1 - D )2 L
with VL > 1.4 V . Example: Increase the turn on voltage of a VIN = 3.3V boost converter from 1.4V to 2.75V. Using VH = 2.75V and R4 = 100K in (12),
R3 = 150K .
(14)
The resulting UVLO hysteresis is:
VHYS = IHYSR3 = 4.6A * 150K = 0.69V .
z2 decreases with increasing duty cycle D and increasing IOUT. Using the 5V to 12V boost regulator (1.35MHz) in Figure 1(a) as an example,
ROUT
5V = 6.8 0.74 A
I OUT VOUT ESR R1 C2 R OUT
V IN
POWER STAGE C5
COMP R3 C6 C4 RO
Gm
+
FB
1.242V VOLTAGE REFERENCE
R2
Figure 8. Simplified Block Diagram of a Boost Converter
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SC4501
POWER MANAGEMENT Application Information
5 12 + 0.5 = 0.62 D= 0 .3 1- 12 + 0.5 1-
p1 = -
1 1 =- RO C 4 4.7M * 820pF
= -260 rads -1 = -41Hz
Therefore
p 2 2 = 29.4Krads-1 = 4.68KHz (6.8 ) * (10F )
6.8 * (1 - 0.62)2 = 209 Krads -1 = 33.3KHz 4.7H
C4 and R3 also forms a zero with angular frequency:
Z1 = -
1 1 =- 30.9K * 820pF R 3C 4
and
Z 2
= -39.5 Krads -1 = -6.3 KHz
The poles p1, p2 and the RHP zero z2 all increase phase shift in the loop response. For stable operation, the overall loop gain should cross 0dB with -20dB/decade slope. Due to the presence of the RHP zero, the 0dB crossover frequency
The spacing between p2 and z2 is the closest when the converter is delivering the maximum output current from the lowest VIN. This represents the worst-case compensation condition. Ignoring C5 and C6 for the moment, C4 forms a low frequency pole with the equivalent output resistance RO of the error amplifier:
Amplifier Open Loop Gain 49dB RO = = = 4.7M Transconduc tan ce 60 -1
z2 . Placing z1 near p2 nulls its 3 effect and maximizes loop bandwidth. Thus
should not be higher than
R 3C 4 VOUT C2 2IOUT (MAX )
(15)
R3 determines the mid-band loop gain of the converter. Increasing R3 increases the mid-band gain and the crossover
GND
C3 R3 C4 C6 R2 U1 C1 R4
SHDN
R1
C5 C2 D1
L1
VOUT
VIN
Figure 9. Suggested PCB Layout for the SC4501. Notice that there is no via directly under the device. All vias are 12mil in diameter.
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SC4501
POWER MANAGEMENT Application Information
frequency. However it reduces the phase margin. The values of R3 and C4 can be determined empirically by observing the inductor current and the output voltage during load transient. Compensation is optimized when the largest R3 and the smallest C4without producing ringing or excessive overshoot in its inductor current and output voltage are found. Figures 10(b), 11(c), 12(b) and 12(c) show load transient responses of empirically optimized DC-DC converters. In a battery-operated system, compensating for the minimum VIN and the maximum load step will ensure stable operation over the entire input voltage range. C5 adds a feedforward zero to the loop response. In some cases it improves the transient speed of the converter. C6 rolls off the gain at high frequency. This helps to stabilize the loop. C5 and C6 are often not needed. Board Layout Considerations In a step-up switching regulator, the output filter capacitor, the main power switch and the rectifying diode carry switched currents with high di/dt. For jitter-free operation, the size of the loop formed by these components should be minimized. Since the power switch is integrated inside the SC4501, grounding the output filter capacitor next to the SC4501 ground pin minimizes size of the high di/dt current loop. The input bypass capacitors should also be placed close to the input pins. Shortening the trace at the SW node reduces the parasitic trace inductance. This not only reduces EMI but also decreases the sizes of the switching voltage spikes and glitches. Figure 9 shows how various external components are placed around the SC4501. The frequency-setting resistor should be placed near the ROSC pin with a short ground trace on the PC board. These precautions reduce switching noise pickup at the ROSC pin. To achieve a junction to ambient thermal resistance (JA) of 40C/W, the exposed pad of the SC4501 should be properly soldered to a large ground plane. Use only 12mil diameter vias in the ground plane if necessary. Avoid using larger vias under the device. Molten solder may seep through large vias during reflow, resulting in poor adhesion, poor thermal conductivity and low reliability.
Typical Application Circuits
VIN 3.3V 6 OFF ON 3 C1 2.2F C3 47nF 8 IN SHDN
L1 3.3H 5 SW FB COMP ROSC 7 R4 9.31K 4 2 1 SC4501 SS GND
D1
VOUT 12V, 0.4A R1 174K
10BQ015
C2 10F R3 22.1K C4 1.5nF R2 20K
40s/div L1: Cooper-Bussmann SD25-3R3 Figure 10(a). 1.35 MHz All Ceramic Capacitor 3.3V to 12V Boost Converter. Pinout Shown is for MSOP-8 Upper Trace : Output Voltage, AC Coupled, 1V/div Lower Trace : Inductor Current, 0.5A/div Figure 10(b). Load Transient Response of the Circuit in Figure 10(a). ILOAD is switched between 0.1A and 0.4A at 1A/s.
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SC4501
POWER MANAGEMENT Typical Application Circuits
Efficiency
95
2.6 - 4.2V
L1 1.8H 6 OFF ON 3 IN SHDN SC4501 8 SS GND C3 47nF 4 COMP ROSC 7 R4 10.7K 1 5 SW FB 2
D1
VOUT = 5V
VOUT 5V, 0.8A
Efficiency (%)
90 85 80 75 70 65
1.2MHz VIN = 4.2V
10BQ015 R1 301K
1-CELL LI-ION
C1 2.2F
C2 10F R3 17.4K C4 1nF R2 100K
60 VIN = 3.6V 55 50 0.001 VIN = 2.6V
0.010
0.100
1.000
L1: Sumida CR43 Figure 11(a). 1.2 MHz All Ceramic Capacitor Single Li-ion Cell to 5V Boost Converter.
Load Current (A)
Figure 11(b). Efficiency of the Single Li-ion Cell to 5V Boost Converter in Figure 11(a).
VIN=2.6V
40s/div Upper Trace : Output Voltage, AC Coupled, 0.5V/div Lower Trace : Inductor Current, 0.5A/div Figure 11(c). Load Transient Response of the Circuit in Figure 11(a). ILOAD is switched between 0.2A and 0.7A at 1A/s.
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SC4501
POWER MANAGEMENT Typical Application Circuits
4-CELL 3.6 - 6V
L1 4.9H 6 OFF ON 3 C1 2.2F 8 IN SHDN SC4501 SS GND C3 47nF 4 COMP ROSC 7 R4 7.68K 1 5 SW FB 2
C6
D1
VOUT 5V
2.2F
10BQ015 C5 47pF R1 60.4K
C2 10F R3 20K C4 560pF L2 4.9H R2 20K
L1 and L2: Coiltronics CTX5-1 Figure 12(a). 1.5 MHz All Ceramic Capacitor 4-Cell to 5V SEPIC Converter. Pinout Shown is for MSOP-8.
VIN=3.6V
VIN=6V
40s/div Upper Trace : Output Voltage, AC Coupled, 0.2V/div Lower Trace : Input Inductor Current, 0.2A/div Figure 12(b). Load Transient Response of the Circuit in Figure 12(a). ILOAD is switched between 50mA and 350mA at 1A/s.
40s/div Upper Trace : Output Voltage, AC Coupled, 0.2V/div Lower Trace : Input Inductor Current, 0.2A/div Figure 12(c). Load Transient Response of the Circuit in Figure 12(a). ILOAD is switched between 80mA and 600mA at 1A/s.
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SC4501
POWER MANAGEMENT Typical Application Circuits
D2 D3 D4 D5 OUT2 23V (10mA) C8 1F
C5 0.1F
C6 0.1F
C7 0.1F
3.3V
L1 2.2H R5 150K 3 6 IN SHDN SC4501 8 R6 100K C3 47nF SS GND 4 COMP ROSC 7 R4 7.68K 1 R3 40.2K C4 820pF 5 SW FB 2
D1
OUT1 8V (0.55A) R1 274K
10BQ015
C1 2.2F
C9 0.1F R2 49.9K
C2 10F
D7
OUT3 -8V (10mA) C10 1F
L1 : Cooper-Bussmann SD25-2R2 D2 - D7 : BAT54S
D6
Figure 13(a). 1.5MHz Triple-Output TFT Power Supply.
CH4
CH4
CH1
CH1
CH2
CH2
CH3
CH3
4ms/div CH1 : OUT1 Voltage, 5V/div CH2 : OUT2 Voltage, 10V/div CH3 : OUT3 Voltage, 5V/div CH4 : Input Voltage, 2V/div Figure 13(b). TFT Power Supply VIN Start-up Transient.
2ms/div CH1 : OUT1 Voltage, 5V/div CH2 : OUT2 Voltage, 10V/div CH3 : OUT3 Voltage, 5V/div CH4 : SHDN Voltage, 2V/div Figure 13(c). TFT Power Supply Start-up Transient as the SHDN Pin is stepped from 0 to 2V.
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SC4501
POWER MANAGEMENT Typical Application Circuits
- 3.4V to 3.8V + 0.7A (FLASH) 0.2A (TORCH) R6 0.1 R1 698 D2 LXCL-PWF1 D1
2.6 - 4.2V
L1 2.2H SUMIDA CR43 +
10BQ015 + 1/2 LM358
1-CELL LI-ION C1 2.2F
6 OFF ON 3 IN SHDN SC4501 8 SS GND C3 10nF 4
5 SW FB COMP ROSC 7 R4 8.06K C4 10nF R5 10K 2 1 C5 0.1F R6 17.4K
Q1 MMBT3904T
C2 4.7F R2 43.2K
M1 MMBF2201NT1
TORCH FLASH
Figure 14(a). 1.4MHz LuxeonTM Flash White LED Driver for Camera Phones
V IN = 2.6V CH1 CH2
VIN = 4.2V CH1 CH2
CH3
CH3
CH4
CH4
4ms/div (b) CH1 : Torch/Flash Control Voltage, 5V/div CH2 : FB Pin Voltage, 1V/div CH3 : LED Current, 0.5A/div CH4 : Inductor Current, 1A/div
4ms/div (c)
Figure 14(b) and 14(c). Photo Flash LED Current is Switched Between Torch Mode (0.2A) and Flash Mode (0.7A). Higher LED Current (>0.7A) in Flash Mode is Possible with Fresh Battery.
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SC4501
POWER MANAGEMENT Outline Drawing - MSOP-8L-EDP
e/2 A N 2X E/2 E1 PIN 1 INDICATOR ccc C 2X N/2 TIPS 12 e B aaa C SEATING PLANE C D A2 A A1 bxN bbb F EXPOSED PAD 0.25 F DETAIL (L1) C A-B D GAGE PLANE L c E D
DIMENSIONS INCHES MILLIMETERS DIM MIN NOM MAX MIN NOM MAX
A A1 A2 b c D E1 E e F L L1 N 01 aaa bbb ccc .043 .006 .000 .037 .030 .009 .015 .009 .003 .114 .118 .122 .114 .118 .122 .193 BSC .026 BSC .068 .076 .080 .016 .024 .032 (.037) 8 0 8 .004 .005 .010 1.10 0.15 0.00 0.75 0.95 0.22 0.38 0.08 0.23 2.90 3.00 3.10 2.90 3.00 3.10 4.90 BSC 0.65 BSC 1.73 1.93 2.03 0.40 0.60 0.80 (0.95) 8 8 0 0.10 0.13 0.25
H
01
A
BOTTOM VIEW SIDE VIEW
NOTES: 1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). -HTO BE DETERMINED AT DATUM PLANE 2. DATUMS -A- AND -B-
SEE DETAIL
A
3. DIMENSIONS "E1" AND "D" DO NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. 4. REFERENCE JEDEC STD MO-187, VARIATION AA-T.
Land Pattern - MSOP-8L-EDP
F
DIM
(C) G F Z C F G P X Y Z
DIMENSIONS INCHES MILLIMETERS
(.161) .081 .098 .026 .016 .063 .224 (4.10) 2.08 2.50 0.65 0.40 1.60 5.70
P X
NOTES: 1.
THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY. CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR COMPANY'S MANUFACTURING GUIDELINES ARE MET.
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SC4501
POWER MANAGEMENT Outline Drawing - MLPD-10, 3 x 3mm
A E B
DIM
A A1 A2 b C D E e L N aaa bbb
DIMENSIONS INCHES MILLIMETERS MIN NOM MAX MIN NOM MAX
.031 .039 .000 .002 (.008) .007 .009 .011 .074 .079 .083 .042 .048 .052 .114 .118 .122 .020 BSC .012 .016 .020 10 .003 .004 0.80 1.00 0.00 0.05 (0.20) 0.18 0.23 0.30 1.87 2.02 2.12 1.06 1.21 1.31 2.90 3.00 3.10 0.50 BSC 0.30 0.40 0.50 10 0.08 0.10
E PIN 1 INDICATOR (LASER MARK)
A aaa C A1 C 1 LxN 2 A2 C SEATING PLANE
D
N e bxN bbb CAB
NOTES:
1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES). 2. COPLANARITY APPLIES TO THE EXPOSED PAD AS WELL AS TERMINALS.
Land Pattern - MLPD-10, 3 x 3mm
K
DIM
H C G H K P X Y Z
DIMENSIONS INCHES MILLIMETERS
(.112) .075 .055 .087 .020 .012 .037 .150 (2.85) 1.90 1.40 2.20 0.50 0.30 0.95 3.80
(C)
G
Z
Y X P
NOTES: 1. THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY. CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR COMPANY'S MANUFACTURING GUIDELINES ARE MET.
Contact Information
Semtech Corporation Power Management Products Division 200 Flynn Road, Camarillo, CA 93012 Phone: (805)498-2111 FAX (805)498-3804
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